Sensing circuit, corresponding amplifier, apparatus and method

ABSTRACT

A switching amplifier, such as a Class D amplifier, includes a current sensing circuit. The current sensing circuit is formed by replica loop circuits that are selectively coupled to corresponding output inverter stages of the switching amplifier. The replica loop circuits operated to produce respective replica currents of the output currents generated by the output inverter stages. A sensing circuitry is coupled to receive the replica currents from the replica loop circuits and operates to produce an output sensing signal as a function of the respective replica currents.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No. 15/984,942 filed May 21, 2018, which claims the priority benefit of Italian Application for Patent No. 102017000058135, filed on May 29, 2017, the disclosures of which are hereby incorporated by reference in their entireties to the maximum extent allowable by law.

TECHNICAL FIELD

The description relates to current sensing in amplifiers.

One or more embodiments may provide a high-precision current sense for switching amplifiers, suitable for a full integration in the amplifier circuit.

BACKGROUND

Various applications involve sensing (for example measuring) the filtered output current of a switching amplifier, for example the output current provided by an amplifier to a load such as a resistive load through a filter such as an external low-pass for example LC filter.

A Class D amplifier may be exemplary of such a switching amplifier.

Despite the extensive activity in that area, improved solutions are desirable in respect of various aspects such as, for example:

-   -   avoiding the possible presence of external components such as an         external sense resistance with the associated efficiency loss,     -   effectively addressing technological issues as possibly         associated to a load terminal reaching levels above the bridge         supply voltage of the amplifier or below ground level,     -   the possibility of supplying a sense circuit and the output         bridges of the amplifier with a same voltage,     -   possibly filtering the ripple in the bridge current, for example         by sensing its mean value, that is the load current, and     -   achieving high accuracy and precision, for example with offset         cancellation.

There is a need in the art to contribute in providing such an improved solution.

SUMMARY

One or more embodiments relate to an amplifier (for example a Class D amplifier), a corresponding apparatus (for example, a micromirror application wherein sensing the output current from a Class D amplifier makes it possible to sense spurious current components due to the physical behavior of the mirror) and a corresponding method.

The claims are an integral part of the disclosure of embodiments as provided herein.

One or more embodiments may sense (directly) the current flowing in the output bridges of a switching amplifier such as for example a Class D amplifier.

One or more embodiments may sense (scaled) currents in the (for example NMOS transistor-based) output inverters of the amplifier through a replica loop across each device.

In one or more embodiments, the (for example scaled) currents may be converted into a voltage signal with such a voltage signal possibly buffered to drive an auxiliary circuit for measurement purposes.

In one or more embodiments, high precision is facilitated by using an offset cancellation procedure combined with a sample-and-hold mechanism providing for the cancellation of the ripple which may be present on the bridge current.

In one or more embodiments, a bridge mean current can be sensed which is indicative of the load current.

In one or more embodiments high accuracy is facilitated by resorting to an amplifier auto-zero offset technique.

One or more embodiments may provide cancellation of superposed ripple due to operation of the switching amplifier, thus facilitating accurate sensing of the mean load current.

One or more embodiments facilitate avoiding using any external components such as an external sensing resistor thus increasing driving efficiency.

In an embodiment, a circuit comprises: replica loop circuits couplable to output inverter stages of a switching amplifier producing amplifier output currents, the replica loop circuits configured for producing respective replica currents of the output currents, and a sensing circuit coupled to the replica loop circuits to receive therefrom the respective output current replicas, the sensing circuit configured to produce an output sensing signal as a function of the respective output current replicas from the replica loop circuits.

The sensing circuit may include a current-to-voltage converter circuit configured for converting the respective output current replicas to a voltage output sensing signal. The sensing circuit may further include an output buffer configured for buffering the voltage signal and producing therefrom a buffered sensing signal.

The replica loop circuits may include current mirror circuits providing replicas of the output currents. The current mirror circuits may include a down-scaling current mirror circuit to provide a respective scaled-down replica of the output current.

Where the output inverter stages generate switched output currents having relatively higher and smaller duty-cycles, the replica loop circuits are selectively activated to act on the switched output current from the inverter stages having a smaller duty-cycle.

The replica loop circuit may include a loop amplifier having associated auto-zero circuitry.

A sampling circuit is activated to sample the output sensing signal at timed instants synchronized with a clock signal that is clocking operation of the switching amplifier. Activation of the sampling circuit occurs responsive to a sampling timing signal having rising and falling edges, where the falling edges of the sampling timing signal are synchronized with the rising edges of the clock signal clocking operation of the switching amplifier. The sampled output sensing signal is indicative of a mean value of the output current from the amplifier.

In an embodiment, a method comprises: providing a switching amplifier having output inverter stages producing amplifier output currents, coupling to the output inverter stages of the switching amplifier replica loop circuits producing respective replicas of the output currents, coupling to the replica loop circuits at least one sensing circuit receiving from the replica loop circuits respective output current replicas, and producing via the sensing circuit an output sensing signal as a function of the respective output current replicas.

BRIEF DESCRIPTION OF THE DRAWINGS

One or more embodiments will now be described, by way of example only, with reference to the annexed Figures wherein:

FIG. 1 is a block diagram illustrative of a possible context of use of embodiments,

FIG. 2 is a circuit diagram exemplary of embodiments,

FIGS. 3 and 4 are diagrams exemplary of possible operation of the circuit of FIG. 2,

FIG. 5 is further exemplary of the principles of operation exemplified in FIGS. 3 and 4,

FIG. 6 is a simplified circuit diagram exemplary of possible operation of embodiments,

FIGS. 7A through 7D are exemplary of possible timing in operation of embodiments,

FIGS. 8A through 8D are exemplary of another circuit representation exemplary of possible operation of embodiments,

FIG. 9 is a circuit diagram exemplary of possible features of embodiments,

FIGS. 10A through 10E are exemplary of possible timing in operation of embodiments,

FIG. 11 is a circuit diagram exemplary of possible features of embodiments, and

FIGS. 12A through 12E are exemplary of possible timing in operation of embodiments.

DETAILED DESCRIPTION

In the ensuing description, one or more specific details are illustrated, aimed at providing an in-depth understanding of examples of embodiments of this description. The embodiments may be obtained without one or more of the specific details, or with other methods, components, materials, etc. In other cases, known structures, materials, or operations are not illustrated or described in detail so that certain aspects of embodiments will not be obscured.

Reference to “an embodiment” or “one embodiment” in the framework of the present description is intended to indicate that a particular configuration, structure, or characteristic described in relation to the embodiment is comprised in at least one embodiment. Hence, phrases such as “in an embodiment” or “in one embodiment” that may be present in one or more points of the present description do not necessarily refer to one and the same embodiment. Moreover, particular conformations, structures, or characteristics may be combined in any adequate way in one or more embodiments.

The references used herein are provided merely for convenience and hence do not define the extent of protection or the scope of the embodiments.

In FIG. 1, reference 100 indicates as a whole an arrangement including a switching amplifier 10 including outputs OUT_(H) (“high”) and OUT_(L) (“low”) supplying load current to a load R_(LOAD) such as for example a resistive load.

The amplifier 10 may include, for example, a Class D Amplifier with an associated output filter (for example a “pi” low-pass filter) for example including two inductors L_(F) and a capacitor C_(F) with the inductors L_(F) coupled at their “proximal” ends to the amplifier stage 10 to be traversed by the current from the outputs OUT_(H), OUT_(L) and the capacitor C_(F) coupled across the “distal” terminals of the inductors L_(F) with the (low-pass) filtered signal applied to the load R_(LOAD).

The principles underlying operation of such a switching amplifier arrangement are known in the art, thus making it unnecessary to provide a more detailed description herein.

As noted, a Class D amplifier with quaternary modulation may be exemplary of such an application. A possible scenario of use of such an arrangement is a micromirror application wherein sensing the output current from the Class D amplifier makes it possible to sense spurious current components due to the physical behavior of the mirror. These currents may thus be compensated by means of a cancelling process to increase mirror driving accuracy. Reference to that possible area of application is merely exemplary and not meant to be limiting of the scope of embodiments.

In various applications, operation of the amplifier 10 will involve sensing (for example measuring) the filtered output signal (current) from the amplifier 10 as exemplary of the current supplied to the load R_(LOAD) through the external (for example LC) filter.

FIG. 1 is exemplary of performing such a sensing action via a sensing resistor R_(SENSE) arranged in series with the load R_(LOAD), with the voltage drop V_(SENSEH)−V_(SENSEL) across the resistor R_(SENSE) indicative of the magnitude of the current supplied to the load R_(LOAD).

As noted, such an arrangement may undesirably involve the use of an external component such as the resistor R_(SENSE) with an associated efficiency loss.

A general exemplary layout of one or more embodiments is shown in FIG. 2, where 10H and 10L indicate the output inverter stages of the switching amplifier 10 (not visible as a whole in FIG. 2).

The output nodes OUT_(H), OUT_(L) of the inverter stages 10H, 10L provide output currents to be supplied to the load R_(LOAD) via the filter L_(F), C_(F).

One or more embodiments may include replica loop circuits 12H, 12L to provide replicas (for example scaled-down replicas) I_(DMYH), I_(DMYL) of the currents supplied towards the load R_(LOAD) via the output nodes OUT_(H), OUT_(L) of the amplifier 10.

In one or more embodiments, the replica loop circuits 12H, 12L may be integrated in the amplifier 10.

Reference 14 in FIG. 2 indicates a current-to-voltage converter sensitive to the replica currents I_(DMYH), I_(DMYL) and configured for generating a corresponding output (voltage) signal V_(C) which may be fed to an output buffer circuit 16 to produce an output sensing signal (for example a voltage signal) V_(SENSE).

In one or more embodiments the converter circuit 14 and/or the output buffer 16 may be integrated in the amplifier 10 as previously indicated for the replica loop circuits 12H, 12L.

In one or more embodiments, operation of the replica loop circuits 12H, 12L may be controlled (as schematically indicated by switches 20H, 20L in FIG. 2) in such a way that the replica loop circuits 12H, 12L may be rendered active on the switching (for example Class D) output inverter 10H, 10L having the smaller duty-cycle as conventionally contemplated for Class D operation.

This type of operation is exemplified in FIGS. 3 and 4 with reference to situations where:

-   -   the duty-cycle at output node OUT_(L) is higher than the         duty-cycle at output node OUT_(H) (FIG. 3), and     -   the duty-cycle at output node OUT_(L) is lower than the         duty-cycle at output node OUT_(H) (FIG. 4).

As used herein, “higher” and “lower” are intended to refer to relative duty-cycle values T_(ON)/(T_(ON)+T_(OFF)) of the switched (for example PWM modulated) signals.

In one or more embodiments, the replica loop circuits 12H, 12L may include K:1 current mirrors including two transistors (for example NMOS transistors), namely:

-   -   M_(OUTH) and M_(SH) (this latter providing I_(DMYH)), and     -   M_(OUTL) and M_(SL) (this latter providing I_(DMYL)).

In the situation exemplified in FIG. 3, the load current I_(L) flows from the output node OUT_(L) to the node OUT_(H) through the external LC filter so that (only) M_(OUTH) is traversed by this current which is mirrored (and scaled down) by the replica loop circuit 12H.

The two transistors M_(OUTH) and M_(SH) in the K:1 current mirror have their control electrodes (gates in the case of a field effect transistors such as MOS transistors) in common and the replica loop includes a differential stage 22H which forces their, for example, drain nodes to be equal, thus providing current mirroring and scaling (by a factor K), so that I_(DMYH)=I_(L)/K.

The replica loop 12H being active is exemplified by the switch 20H being represented in a closed (that is conductive) condition.

In an arrangement as exemplified in the figures, the mirrored, scaled-down current flows through a feedback resistor R1 of an amplifier 140 included in the current/voltage converter 14.

In an arrangement as exemplified in the figures the amplifier 140 includes a differential stage receiving a (voltage) reference signal V_(REF) at one of its inputs (for example non-inverting) with the feedback resistor R1 coupled to the replica loop circuit 12H at the other (for example inverting) input.

In an arrangement as exemplified in the figures, an output resistor R2 is set between the output of the differential stage 140 and the buffer stage 16 (and the loop replica circuit 12L).

The current flowing in the feedback resistor of the differential stage 140 of the converter 14 may be buffered at 16 to generate an output signal V_(SENSE)=V_(REF)+I_(DMYH).

Operation in the complementary conditions as exemplified in FIG. 4 is essentially identical save that the two loop replica circuits 12H, 12L “swap” their roles with the loop 12L leading to the generation of a current I_(DMYL)=I_(L)/K which flows in the output resistor R2 of the amplifier 140 in the converter 14 and is buffered to generate a sensing signal V_(SENSE)=V_(REF)−I_(DMYL)*R2/K.

By combining the two sense paths the signal V_(SENSE) provides a good replica of the load current(s) of the switching (for example Class D) amplifier as shown in the right-hand side of FIG. 5 (assuming for example R1=R2=R).

As discussed previously, the replica loops 12H, 12L may operate with the target of facilitating rendering the drains of the transistors M_(OUTH,L) and M_(SH,L) (where M_(OUTH,L) indicates M_(OUTH) and M_(OUTL), while M_(SH,L) indicates M_(SH) and M_(SL)) equal.

However, it was observed that the transistors M_(OUTH,L) may exhibit (very) small values for the “on” resistance R_(ON), this resulting in a correspondingly small value for V_(OUT) even with relatively high load currents.

As a result, a correct operation of the replica loop circuit may be facilitated by the loop amplifiers 22H, 22L having an offset comparable or smaller than V_(OUT).

For that reason, one or more embodiments may adopt an amplifier “auto-zero” technique in order to achieve high accuracy.

FIG. 6 is exemplary of a possible implementation of a replica loop with auto-zero, where the representation of FIG. 6 applies to both replica loop circuits 12H, 12L including the loop amplifiers 22H, 22L.

The chronograms of FIGS. 7A through 7D are exemplary of the timing of operation of switches SWA_A, SWA_B and 24, 26, 28 in four working phases as represented in FIGS. 7A through 7D by a Φ1, Φ2, Φ3 and Φ4.

In the four diagrams of FIGS. 7A through 7D:

-   -   the diagram of FIG. 7A is exemplary of the clock signal used for         switching (Class D modulation),     -   the diagrams in FIG. 7B (dashed line and solid line) are         exemplary of the outputs of the two single-ended Class D nodes,         with the replica branch 12H, 12L active only on the output node         having the smaller duty-cycle as discussed previously in         connection with FIGS. 3 and 4,     -   the diagram in FIG. 7C is exemplary of the enable signal for the         switch SWA_A causing the output voltage from the transistor         M_(OUTH,L) to be sampled on a capacitor CSeH_A, and     -   the diagram in FIG. 7D is exemplary of the enable signal for the         switch SWA_B causing the output voltage from the transistor         M_(OUTH,L) to be sampled on capacitor CSeH_B.

FIGS. 8A through 8D are exemplary of the various enable (switch ON, that is conductive) and non-enable (switch OFF, that is non-conductive) conditions of the various switches presented in FIG. 6 in the four phases Φ1, Φ2, Φ3, Φ4.

Briefly (in the following the suffixes “H” and “L” distinguishing the two replica loop circuits are dropped for simplicity, operation of two circuits being otherwise the same):

Φ1 (FIG. 8A): M_(OUT) is ON→V_(OUT)[n]=R_(ON)*I_(L). V_(OUT)[n] is sampled on C_(SeH_A), whereas the amplifier is in Sensing Phase using V_(OUT)[n−3] (voltage sampled on C_(SeH_B) in Φ2 of the previous clock cycle) as the reference voltage→V_(DMY)=V_(OUT)[n−3]−V_(OS)[n]+V_(OS)[n−1] and since V_(OS)[n]=V_(OS)[n−1]→V_(DMY)=V_(OUT)[n−3]→I_(DMY)=I_(L)/K→I_(DMY) in this phase is directly proportional to I_(L) and can be used to generate V_(SENSE).

Φ2 (FIG. 8B): M_(OUT) is ON→V_(OUT)[n]=R_(ON)*I_(L). V_(OUT)[n] is sampled on C_(SeH_B), whereas the amplifier is in Autozero Phase using V_(OUT)[n−1] (voltage sampled on C_(SeH_A) in Φ1 of the same clock cycle) as the reference voltage→V_(DMY)=V_(OUT)[n−1]−V_(OS)[n]→I_(DMY)=I_(L)/K+V_(OS)[n]/(K*R_(ON)). The term V_(OS)[n]/(K*R_(ON)) represents an error, thus the I_(DMY) current, during this phase, is not useful for generating V_(SENSE).

Φ3 (FIG. 8C): M_(OUT) is OFF Both SWA_A and SWA_B are open. The amplifier is in Autozero Phase using V_(OUT)[n−2] (voltage sampled on C_(seH_A) in Φ1 of the same clock cycle) as the reference voltage→V_(DMY)=V_(OUT)[n−2]−V_(OS)[n]→I_(DMY)=I_(L)/K+V_(OS)[n]/(K*R_(ON)). The term V_(OS)[n]/(K*R_(ON)) represents an error, thus the I_(DMY) current, during this phase, is not useful for generating V_(SENSE).

Φ4 (FIG. 8D): M_(OUT) is OFF Both SWA_A and SWA_B are open. The amplifier is in Sensing Phase using V_(OUT)[n−2] (voltage sampled on C_(SeH_B) in f2 of the same clock cycle) as the reference voltage→V_(DMY)=V_(OUT)[n−2]−V_(OS)[n]+V_(OS)[n−1] and since V_(OS)[n]=V_(OS)[n−1]→V_(DMY)=V_(OUT)[n−2]→I_(DMY)=I_(L)/K→I_(DMY) in this phase is directly proportional to I_(L) and can be used to generate V_(SENSE).

After the replica current I_(DMY) (that is I_(DMYH) or I_(DMYL), respectively, as a function of the active replica loop circuit considered) is converted to voltage, the output from the converter 14 may be sampled during the phase Φ1, that is when this is (directly) proportional to I_(L) and is exempt from error, thanks to offset cancellation thus facilitating buffering at 16 to drive a load.

In one or more embodiments such a sampling action can be implemented in the buffer circuit 16 as schematically illustrated in FIG. 9.

In FIG. 9 reference 160 indicates an input stage receiving the signal Vc from the converter circuit 14 and reference 162 indicates an output stage providing an output sensing signal V_(SENSE).

In FIG. 9 a sampling switch EN_(seH) 164 set between the input and output stages 160, 162 of the buffer circuit 16 is exemplary of a possible implementation of the sampled output buffer with the respective sampling timing signal EN_(S&H) added to the diagrams of FIGS. 7A through 7D as shown in the lower portion of FIGS. 10A through 10E thus showing that the replica current may be effectively sampled at the end of the phase Φ1.

The foregoing discussion assumes that the mean value I_(L) of the load current (for example a sinusoidal current if the input to the switching amplifier for example Class D input) is a sine wave.

In one or more embodiments, the load current designated I_(BRIDGE) may have superimposed a ripple due to the way of working of a Class D amplifier.

An accurate sensing of the mean (average) load current is facilitated by such a current ripple being cancelled.

In one or more embodiments, deriving the falling edge of the sampling signal EN_(S&H) signal from the rising edge of the clock used for Class D modulation (see for example the diagrams a) in FIGS. 7A through 7D and FIG. 9) may facilitate having the voltage sample at the end of EN_(S&H) to correspond (exactly) to the mean load current I_(L).

In that way, the possibility exists of automatically filtering the ripple of the bridge current as exemplified in the diagrams of the FIGS. 12A through 12E designate:

FIG. 12A: the clock signal used for Class D modulation V_(CK),

FIGS. 12B and 12C: the outputs form the two output nodes of the Class D amplifier OUT_(H), OUT_(L),

FIG. 12D: the sampling signal EN_(S&H), and

FIG. 12E: a possible “sawtooth” behavior (deliberately emphasized for ease of understanding) of the ripple I_(BRIDGE) with respect to the average value I_(L).

FIG. 12E exemplifies how sampling taking place as discussed previously makes it possible to automatically filter out such a ripple.

A circuit arrangement according to one or more embodiments may include:

-   -   replica loop circuits (for example 12H, 12L) couplable to output         inverter stages (for example 10H, 10L) of a switching amplifier         (for example 10) producing amplifier output currents, the         replica loop circuits configured for producing respective         replicas (for example I_(DMYH), I_(DMYL)) of the output currents         from the output inverter stages, and     -   at least one sensing circuit (for example 14, 16) coupled to the         replica loop circuits to receive therefrom the respective output         current replicas, the sensing circuit configured to produce an         output sensing signal (for example V_(SENSE)) as a function of         the respective output current replicas from the replica loop         circuits.

In one or more embodiments the at least one sensing circuit may include a current-to-voltage converter circuit (for example 14) configured for converting the respective output current replicas to a voltage output sensing signal.

In one or more embodiments the at least one sensing circuit may include an output buffer (for example 16) configured for buffering the voltage signal and producing therefrom a buffered sensing signal (for example V_(SENSE)).

In one or more embodiments, the replica loop circuits may include current mirrors (for example M_(OUTH), M_(SH); M_(OUTL), M_(SL)) providing replicas of the output currents.

In one or more embodiments, the current mirrors may include down-scaling current mirrors provide respective scaled-down replicas of the output currents.

In one or more embodiments, with the output inverter stages generating switched output currents having relatively higher and smaller duty-cycles, the replica loop circuits may be selectively activatable (for example 20H, 20L) to act on the switched output current from the inverter stages having a smaller duty-cycle.

In one or more embodiments, the replica loop circuits may include loop amplifiers (for example 22H, 22L) having associated auto-zero circuitry (for example SWA_A, SWA_B, CSeH_A, CSeH_B, 24, 26, 28).

One or more embodiments may include a sampling circuit (for example 164) activatable to sample the output sensing signal at timed instants synchronized (for example EN_(S&H)) with a clock signal (for example V_(CK)) clocking operation of the switching amplifier.

One or more embodiments may include a sampling circuit activatable by a sampling timing signal having rising and falling edges, the falling edges of the sampling timing signal synchronized with the rising edges of the clock signal clocking operation of the switching amplifier (10), whereby the sampled output sensing signal is indicative of the mean value of the output current from the amplifier.

One or more embodiments may include a switching amplifier (for example a Class D amplifier) including output inverter stages producing amplifier output currents, the amplifier including a circuit arrangement according to one or more embodiments.

Apparatus according to one or more embodiments may include a switching amplifier according to one or more embodiments.

A method according to one or more embodiments may include:

-   -   providing a switching amplifier having output inverter stages         producing amplifier output currents,     -   coupling to the output inverter stages of the switching         amplifier replica loop circuits producing respective replicas of         the output currents, and     -   coupling to the replica loop circuits at least one sensing         circuit receiving from the replica loop circuits respective         output current replicas,     -   producing via the sensing circuit an output sensing signal as a         function of the respective output current replicas.

Without prejudice to the underlying principles, the details and embodiments may vary, even significantly, with respect to what has been described by way of example only, without departing from the extent of protection.

The extent of protection is defined by the annexed claims. 

The invention claimed is:
 1. A circuit, comprising: a current mirror comprising an output drive transistor having a source node and a drain node and a current sensing transistor having a source node and a drain node; a differential amplifier having a first input coupled to the drain node of the output drive transistor and a second input coupled to the drain node of the current sensing transistor; a further transistor having a gate node coupled to an output of the differential amplifier, a source node coupled to the drain node of the current sensing transistor and a drain node generating a replica current output; and an auto-zero circuit, comprising: a first switched capacitor circuit path coupled between the drain node of the output drive transistor and the first input of the differential amplifier; and a second switched capacitor circuit path coupled between the drain node of the output drive transistor and the first input of the differential amplifier.
 2. The circuit of claim 1, further comprising: a control of the first switched capacitor circuit path in a first operational phase to sample a voltage at the drain of the output drive transistor; a control of the first switched capacitor circuit path in a second operational phase to apply the sampled voltage to the first input of the differential amplifier.
 3. The circuit of claim 2, further comprising a control to turn on the output drive transistor during the first and second operational phases.
 4. The circuit of claim 2, wherein the auto-zero circuit further comprises: a capacitor having a first terminal and a second terminal, wherein the second terminal is coupled to the second input of the differential amplifier; a first switch coupled between the first input of the differential amplifier and the first terminal of the capacitor; a second switch coupled between the second terminal and the drain node of the current sensing transistor; and a third switch coupled between the first terminal and the drain node of the current sensing transistor.
 5. The circuit of claim 4, further comprising: a control of the first switch to close during the first operational phase; a control of the second switch to close during the first operational phase; and a control of the third switch to open during the first operational phase.
 6. The circuit of claim 4, further comprising: a control of the first switch to open during the second operational phase; a control of the second switch to open during the second operational phase; and a control of the third switch to close during the second operational phase.
 7. The circuit of claim 2, further comprising: a control of the second switched capacitor circuit path in the first operational phase to apply a previously sampled voltage at the drain of the output drive transistor to the first input of the differential amplifier; and a control of the second switched capacitor circuit path in the second operational phase to sample a voltage at the drain of the output drive transistor.
 8. The circuit of claim 7, further comprising a control to turn on the output drive transistor during the first and second operational phases.
 9. The circuit of claim 7, wherein the auto-zero circuit further comprises: a capacitor having a first terminal and a second terminal, wherein the second terminal is coupled to the second input of the differential amplifier; a first switch coupled between the first input of the differential amplifier and the first terminal of the capacitor; a second switch coupled between the second terminal and the drain node of the current sensing transistor; and a third switch coupled between the first terminal and the drain node of the current sensing transistor.
 10. The circuit of claim 9, further comprising: a control of the first switch to close during the first operational phase; a control of the second switch to close during the first operational phase; and a control of the third switch to open during the first operational phase.
 11. The circuit of claim 9, further comprising: a control of the first switch to open during the second operational phase; a control of the second switch to open during the second operational phase; and a control of the third switch to close during the second operational phase.
 12. The circuit of claim 2, further comprising: a control of the first switched capacitor circuit path in a third operational phase to apply the sampled voltage to the first input of the differential amplifier; and a control of the first switched capacitor circuit path in a fourth operational phase to isolate a capacitance of the first switched capacitor circuit path from both the drain of the output drive transistor and the first input of the differential amplifier.
 13. The circuit of claim 12, further comprising a control to turn off the output drive transistor during the third and fourth operational phases.
 14. The circuit of claim 12, wherein the auto-zero circuit further comprises: a capacitor having a first terminal and a second terminal, wherein the second terminal is coupled to the second input of the differential amplifier; a first switch coupled between the first input of the differential amplifier and the first terminal of the capacitor; a second switch coupled between the second terminal and the drain node of the current sensing transistor; and a third switch coupled between the first terminal and the drain node of the current sensing transistor.
 15. The circuit of claim 14, further comprising: a control of the first switch to close during the fourth operational phase; a control of the second switch to close during the fourth operational phase; and a control of the third switch to open during the fourth operational phase.
 16. The circuit of claim 14, further comprising: a control of the first switch to open during the third operational phase; a control of the second switch to open during the third operational phase; and a control of the third switch to close during the third operational phase.
 17. The circuit of claim 12, further comprising: a control of the second switched capacitor circuit path in the third operational phase to isolate a capacitance of the second switched capacitor circuit path from both the drain of the output drive transistor and the first input of the differential amplifier; and a control of the second switched capacitor circuit path in the fourth operational phase to apply a previously sampled voltage at the drain of the output drive transistor to the first input of the differential amplifier.
 18. The circuit of claim 17, further comprising a control to turn off the output drive transistor during the third and fourth operational phases.
 19. The circuit of claim 17, wherein the auto-zero circuit further comprises: a capacitor having a first terminal and a second terminal, wherein the second terminal is coupled to the second input of the differential amplifier; a first switch coupled between the first input of the differential amplifier and the first terminal of the capacitor; a second switch coupled between the second terminal and the drain node of the current sensing transistor; and a third switch coupled between the first terminal and the drain node of the current sensing transistor.
 20. The circuit of claim 19, further comprising: a control of the first switch to close during the fourth operational phase; a control of the second switch to close during the fourth operational phase; and a control of the third switch to open during the fourth operational phase.
 21. The circuit of claim 19, further comprising: a control of the first switch to open during the third operational phase; a control of the second switch to open during the third operational phase; and a control of the third switch to close during the third operational phase.
 22. A circuit, comprising: a current mirror comprising an output drive transistor having a source node and a drain node and a current sensing transistor having a source node and a drain node; a differential amplifier having a first input coupled to the drain node of the output drive transistor and a second input coupled to the drain node of the current sensing transistor; a further transistor having a gate node coupled to an output of the differential amplifier, a source node coupled to the drain node of the current sensing transistor and a drain node generating a replica current output; and an auto-zero circuit, comprising: a capacitor having a first terminal and a second terminal, wherein the second terminal is coupled to the second input of the differential amplifier; a first switch coupled between the first input of the differential amplifier and the first terminal of the capacitor; a second switch coupled between the second terminal and the drain node of the current sensing transistor; and a third switch coupled between the first terminal and the drain node of the current sensing transistor.
 23. The circuit of claim 22, further comprising: a control of the first switch to close during a first operational phase; a control of the second switch to close during the first operational phase; and a control of the third switch to open during the first operational phase.
 24. The circuit of claim 23, further comprising a control to turn on the output drive transistor during the first operational phase.
 25. The circuit of claim 23, further comprising: a control of the first switch to open during a second operational phase; a control of the second switch to open during the second operational phase; and a control of the third switch to close during the second operational phase.
 26. The circuit of claim 25, further comprising a control to turn on the output drive transistor during the first and second operational phases.
 27. The circuit of claim 22, further comprising: a control of the first switch to open during a third operational phase; a control of the second switch to open during the third operational phase; and a control of the third switch to close during the first operational phase.
 28. The circuit of claim 27, further comprising a control to turn off the output drive transistor during the third operational phase.
 29. The circuit of claim 27, further comprising: a control of the first switch to close during a fourth operational phase; a control of the second switch to close during the fourth operational phase; and a control of the third switch to open during the fourth operational phase.
 30. The circuit of claim 29, further comprising a control to turn off the output drive transistor during the third and fourth operational phases. 